1. Field of the Invention
The present invention relates to charge sensitive amplifiers, and more particularly relates to a charge sensitive amplifier for use on monolithic substrates, which includes a low-noise, active feedback element that exhibits improved linearity and dynamic range.
2. Description of the Prior Art
Charge sensitive amplifiers are used in a variety of commercial, industrial, medical, and scientific instrumentation applications where a signal from a sensor is provided in the form of a small current charge that requires amplification prior to further signal processing or conditioning. These sensors may include low-capacitance silicon detectors, such as those used in particle position sensing, X-ray spectroscopy, and X-ray imaging, as well as sensors used to detect gamma rays emitted in mammography systems, and other low charge output sensing devices.
Charge sensitive amplifiers generally require a high-value feedback resistor to achieve low noise performance. However, high-value resistors are difficult to implement using conventional complimentary metal oxide semiconductor (CMOS) fabrication methods. Circuit configurations have been used in the prior art that include an active element to achieve the desired high resistance in a monolithically formed charge sensitive amplifier. These configurations suffer from several disadvantages, such as periods of inoperability, feedback instability, and very large variations in the resistance of the feedback element due to variations in process, temperature, and power supply voltage.
Charge sensitive amplifiers typically need low-frequency feedback to stabilize the operating point of the amplifier, discharge the feedback capacitor and, when applicable, to absorb leakage current from a corresponding sensor connected to the input of the amplifier. In discrete charge sensitive amplifiers, the feedback network includes a high-value resistor. In monolithic charge sensitive amplifiers, where high-value resistors cannot be integrated, active devices are used.
The major advantage in using active devices in the feedback network is that they are able to adapt to the value of the leakage current. The major drawbacks include non-linearity, noise contributions, and a voltage drop limiting the dynamic range available at the output of the amplifier.
These problems become even more critical as the supply voltage and metal oxide semiconductor field effect transistor (MOSFET) threshold voltage decrease. Conventional charge sensitive amplifiers are affected by at least one of these three major disadvantages.
FIG. 1 is a schematic diagram of a charge sensitive amplifier, which includes a signal amplifier 10 having an input terminal and an output terminal. The input terminal of the signal amplifier 10 is connected to a current supply 12, and a feedback capacitor C1 is connected in parallel across the input and output terminals of the signal amplifier 10.
The circuit also includes an n-channel MOSFET M1 having source, drain, and gate terminals. The drain terminal of MOSFET M1 is connected to the input terminal of the signal amplifier 10 and the source terminal of MOSFET M1 is connected to the output terminal of the signal amplifier 10. The circuit also includes at least one MOSFET M2 and at least one capacitor C2, which operate as a replicable pole-zero cancellation network. The capacitor C2 is connected in series between the output terminal of the signal amplifier 10 and an input terminal of a second signal amplifier 111 in a second amplification stage. A source terminal of MOSFET M2 is connected to the output terminal of the signal amplifier 10 and a drain terminal of MOSFET M2 is connected to the input terminal of the second signal amplifier 11. An impedance 14 is connected in parallel across the input and output terminals of the second signal amplifier 11 to provide feedback.
The configuration shown in FIG. 1 is typically used with CMOS technologies having minimum feature size down to 0.5 μm, which operate at supply voltages of about 3.3 voltages and above. However, this solution becomes substantially more difficult to use with more recent CMOS technologies of 0.25 μm or less, which need to be operated at supply voltages of 2.5 volts or less.
Additional details concerning the configuration shown in FIG. 1 are provided in U.S. Pat. No. 5,793,254; P. O'Connor et al., Ultra Low Noise CMOS Preamplifier-Shaper for X-Ray Spectroscopy, Nuclear Instruments and Methods in Physics Research, A 409, pp. 315–321 (1998); G. De Geronimo et al., A Fully Compensated Continuous Reset System, IEEE Transactions on Nuclear Science, Vol. 47, No. 4, pp. 1458–1462 (2000); and G. Bertuccio et al., MOSFET Diode as a Feedback Reset Element on Charge Amplifiers, IEEE Transactions on Nuclear Science, Vol. 46, No. 3, pp. 757–760 (1999), which are incorporated herein by reference.
MOSFET M2 and capacitor C2 may be replicated N times to provide for a current gain equal to N and the input operating points of amplifiers 10, 11 must be matched. However, an input operating point i1 of the amplifier 10 is about one threshold voltage above ground. The threshold voltage refers to the voltage difference between the gate and source terminals of the signal amplifier 10 required to turn the amplifier on. The output operating point o1, due to the polarity of a direct current (DC) component of an input current I, is thus lower than one threshold voltage of the amplifier 10. In response to a transient current pulse having the same polarity as the current I, the output node o1 must swing negatively and, with low voltage technologies that are characterized by small threshold voltages, the output dynamic range is severely limited in the configuration shown in FIG. 1.
FIG. 2 is a schematic diagram of another conventional charge sensitive amplifier configuration, which includes a signal amplifier 16 having input and output terminals. The circuit includes a capacitor C3 having an input terminal connected to the input terminal of the signal amplifier 16 and a drain terminal of a MOSFET M3. A source terminal of MOSFET M3 is connected to ground and a gate terminal of MOSFET M3 is connected to the gate and drain terminals of a MOSFET M5.
The source terminal of MOSFET M5 is connected to ground and MOSFETs M3, M5 function as a current mirror circuit. The output terminal of the signal amplifier 16 is connected to a gate terminal of a MOSFET M4, and a drain terminal of MOSFET M4 is connected to the drain terminal of MOSFET M5. The source terminal of MOSFET M4 is connected to the remaining terminal of capacitor C3, and a terminal of a resistor R1. The remaining terminal of resistor R1 is connected to a voltage supply VDD.
The second configuration provides a current-to-voltage conversion equal to 1/(ωC3) and the output dynamic range is larger than that of the configuration shown in FIG. 1. A discharge time consent of the feedback capacitor C3 is represented by the product of C3, R1 and n, where n represents a current mirror ratio, which is the ratio of the drain currents for MOSFETs M3, M5. However, the configuration shown in FIG. 2 is non-linear, that is, there is no pole-zero cancellation and the discharge of feedback capacitor C3 depends on the amplitude of the signal. In addition, the configuration shown in FIG. 2 is noisy, that is, the resistor R1 and MOSFETs M3, M4, M5 each contribute substantially to the noise generated by the feedback network.
Further details concerning the configuration shown in FIG. 2 are provided in M. Sampietro et al., Zero-Power Current Conveyor for DC Stabilization and System Reset of Fast Current Pulse Amplifier, IEEE Electronics Letters, Vol. 34, No. 19, pp. 1801–1802 (1998); and M. Sampietro et al., Current Mirror Reset for Low-Power BiCMOS Charge Amplifier, Nuclear Instruments and Methods in Physics Research A 439, pp. 373–377 (2000), which are incorporated herein by reference.
FIG. 3 is a block diagram of a preamplifier and shaper circuit for use with detectors in nuclear spectroscopy. One of the problems with such circuits concerns discharging the preamplifier feedback capacitor C4. For low parallel-noise, high-value resistors in the range of megohms are required. However, with complementary metal oxide semiconductor (CMOS) technology, realistic values can only be provided in the range of tens of kilohms.
To maintain a low discharge current, a current mirror technique using an amplifier, the schematic of which is shown in FIG. 4, is used as a bidirectional current source. By using this method, an equivalent resistor in the order of megohms may be realized in CMOS. The pole-zero cancellation portion 18 of the circuit uses an identical current mirror, which delivers current to the pole-zero circuit equal to the current discharging the preamplifier feedback capacitor C4.
In the preamplifier, a capacitor C6 represents the capacitance of the detector, which is connected across the input of the amplifier 20 and ground. The feedback capacitor C4 is connected in parallel across the input and output terminals of the amplifier 20. A decoupling capacitor C5 is connected in series between an output terminal of the signal amplifier 20 and the pole-zero cancellation network 18.
Additional details concerning the configuration shown in FIG. 3 are provided in R. L. Chase et al., 8-Channel CMOS Preamplifier and Shaper with Adjustable Peaking Time and Automatic Pole-Zero Cancellation, Nuclear Instruments and Methods in Physics Research, A 409, pp. 328–331 (1998); L. Blanquart et al., XPAD, A New Read-out Pixel Chip for X-Ray Imaging, Nuclear Science Symposium Conference Record, IEEE, pp. 92–97, (2000), which are incorporated herein by reference.
The configuration shown in FIG. 3 provides a non-linear current gain equal to about −C5/C4 and the output dynamic range is larger than that of the circuit shown in FIG. 1. The discharge time constant of the feedback capacitor C4 is equal to the product of the values of C4, R2, and N, where N represents the total current scaling factor. However, the configuration shown in FIG. 4 is non-linear, that is, due to the low mirror current and voltage mismatches, pole-zero cancellation is fully effective only when C4 equals C5. In addition, this configuration is noisy, that is, resistor R2 and MOSFETs in the current mirror 22 contribute substantially to the noise generated by the feedback network.
FIG. 5 shows another conventional charge sensitive amplifier, which is configured as a differential feedback amplifier and includes a signal amplifier 24. An input terminal of the signal amplifier 24 is connected to a current supply 26 and a drain terminal of a MOSFET M6. A source terminal of MOSFET M6 is connected to ground and a gate terminal of MOSFET M6 is connected to a second current supply 28.
A drain terminal of a MOSFET M7 is connected to the second current supply 28 and a gate terminal of MOSFET M7 is connected to an output terminal of the signal amplifier 24. A feedback capacitor C7 is connected in parallel across the input and output terminals of the signal amplifier 24. A drain terminal of a MOSFET M8 is connected to the input terminal of the signal amplifier 24 and a source terminal of MOSFET M8 is connected to a voltage supply VDD. A gate terminal of MOSFET M8 and a gate terminal of MOSFET M7 is connected to the voltage supply VDD.
The amplifier shown in FIG. 5 provides a current-to-voltage conversion equal to 1/(ωC7) with an output dynamic range larger than that of the configuration shown in FIG. 1. However, like the configuration shown in FIG. 2, the amplifier in FIG. 5 is non-linear due to the lack of pole-zero cancellation. Discharge of the feedback capacitor C7 depends on the amplitude of the signal. In addition, the configuration shown in FIG. 5 is noisy, that is, each of the MOSFETs M6, M7, M8 and voltage sources 26, 28, 30 contribute substantially to the noise generated by the feedback network.
In addition, this configuration is not self-adaptive to the sensor leakage current, that is, the maximum leakage current that can be absorbed by the circuit is equal to the tail current of the differential pair represented by MOSFETs M7, M8. Leakage currents greater than this value cannot be absorbed and the configuration exhibits excessive noise for detectors exhibiting leakage currents that are smaller than this value.
Further details concerning the amplifier configuration shown in FIG. 5 are provided in F. Krummenacher, Pixel Detectors with Local Intelligence: An IC Designer Point of View, Nuclear Instruments and Methods in Physics Research, A 305, pp. 527–532 (1991); B. Ludewigt et al., A High Rate, Low Noise, X-ray Silicon Strip Detector System, IEEE Transactions on Nuclear Science, Vol. 41, No. 4, pp. 1037–1041 (1994); and P. F. Manfredi et al., The Analog Front-End Section of the BaBar Silicon Vertex Tracker Readout IC, Nuclear Physics B (Proc. Suppl.), Vol. 61B, pp. 532–538 (1998), which are incorporated herein by reference.
Thus, each of the conventional charge sensitive amplifier configurations discussed above suffers from at least one of the three major disadvantages associated with active devices in a feedback network. These disadvantages being non-linearity, noise, and the dynamic range available at the output of the amplifier.